Voltage regulation is commonly required to prevent variation in the supply voltage powering various microelectronic components such as digital ICs, semiconductor memories, display modules, hard disk drives, RF circuitry, microprocessors, digital signal processors and analog ICs, especially in battery powered applications such as cell phones, notebook computers and consumer products.
Since the battery or DC input voltage of a product often must be stepped-up to a higher DC voltage, or stepped-down to a lower DC voltage, such regulators are referred to as DC-to-DC converters. Step-down converters, commonly referred to as “Buck converters,” are used whenever a battery's voltage is greater than the desired load voltage. Step-down converters may comprise inductive switching regulators, capacitive charge pumps, and linear regulators. Conversely, step-up converters, commonly referred to as “boost converters,” are needed whenever a battery's voltage is lower than the voltage needed to power its load. Step-up converters may comprise inductive switching regulators or capacitive charge pumps.
Another type of converter may operate as either a step-up or a step-down converter depending on whether the power input to the converter has a voltage above or below its output voltage. Commonly referred to Buck-boost converters, such circuitry is needed whenever a regulator's input and output are similar in voltage, where variations in the input voltage preclude the use of a simple boost or Buck converter.
One example of such an application requiring both step-up and step-down conversion is supplying a regulated 3.3V output from a lithium ion (Lilon) battery. A Lilon battery exhibits a terminal voltage which decays from 4.2V when fully charged to below 3V when discharged. Since the initial battery voltage is above 3.3V and the final battery voltage is below 3.3V, the converter must be able to step-down initially and step-up later.
Inductive Switching Converters
Of the aforementioned voltage regulators, the inductive switching converter can achieve superior performance over the widest range of currents, input voltages and output voltages. The principles of inductive switching regulator operation are described in detail in application Ser. No. 11/890,818, titled “High-Efficiency DC/DC Voltage Converter Including Down Inductive Switching Pre-Regulator And Capacitive Switching Post-Converter,” which is incorporated herein by reference in its entirety.
Two examples of non-isolated inductive switching regulators, a synchronous Buck step-down converter and synchronous boost step-up converter, are shown in FIGS. 1A and 1B, respectively.
An example of a synchronous Buck converter 1 is shown in FIG. 1A. Converter 1 comprises a power MOSFET 3, an inductor 5, a synchronous rectifier power MOSFET 4 with an intrinsic rectifier diode 8, and a capacitor 6. Operation of MOSFET 3 is controlled by a pulse-width modulation (PWM) controller 2, driving the gate of MOSFET 3. The gate drive may vary in polarity and voltage depending on whether MOSFET 3 is an N-channel or a P-channel MOSFET. Synchronous rectifier MOSFET 4, generally an N-channel MOSFET, is driven out of phase with MOSFET 3, but is not necessarily turned on the entire time when MOSFET 3 is off. In general, MOSFET 4 conducts only during times when diode 8 is conducting.
While the control circuit controlling the converter's operation is referred to as a PWM controller, implying fixed-frequency variable-pulse-width operation, it may alternatively operate in a variable frequency mode where the clock period is allowed to vary, or alternatively alternating between varying and fixed frequency modes depending on load and input conditions.
The energy input from the power source, battery or power input into the DC/DC converter is switched or gated through MOSFET 3. With its positive terminal connected to the battery or input, MOSFET 3 acts like a “high-side” switch controlling the current in inductor 5. Diode 7 is a P-N junction parasitic to MOSFET 3, in parallel to the transistor's drain and source, which remains reverse-biased under normal Buck converter operation. Since diode 7 does not carry current under normal operation, it is illustrated by dotted lines.
By controlling the current in the inductor 5 by controlling the switching and on-time of MOSFET 3, the energy stored in the magnetizing field of inductor 5 can be adjusted dynamically to control the voltage on output filter capacitor 6. The output voltage Vout is therefore fed back to the input of PWM controller 2, which controls the current IL in inductor 5 through the repeated switching of MOSFET 3. The electrical load connected to the converter's output is not shown.
Driven out of phase with MOSFET 3, synchronous rectifier MOSFET 4 conducts some portion of the time when MOSFET 3 is off. With its positive terminal connected to the inductor 5, i.e. to node Vx, and its negative terminal connected the circuit ground, MOSFET 4 acts like a “low-side” switch shunting the current in diode 8. Diode 8 is a P-N junction parasitic to synchronous rectifier MOSFET 4, in parallel to the transistor's drain and source. Diode 8 conducts substantial inductor current only during intervals when both MOSFETs are off.
Both MOSFETs are simultaneously off during every switching transition to prevent shorting the input power source to ground. This so-called “break-before-make” (BBM) interval prevents shoot through conduction by guaranteeing both transistors do not conduct simultaneously and short or “crow-bar” the converter's input and power source.
During this brief BBM interval, diode 8 in parallel to synchronous rectifier MOSFET 4 must, along with any parasitic capacitance associated with diode 8, carry the load current through inductor 5. Unwanted noise can occur during the transitions associated with BBM operation.
If we define the converter's duty factor D as the time that energy flows from the battery or other power source into the DC/DC converter, i.e. during the time that MOSFET switch 3 is on, then the ratio of output to input voltage ratio in a Buck converter 1 is proportionate to its duty factor, i.e.
            V      out              V      in        =      D    ≡                  t        sw            T      where tsw is the time period that MOSFET 3 is turned on during each clock period T.
This relationship for a Buck or synchronous Buck converter is illustrated by curve 17 in FIG. 2A in graph 15. Notice that the Buck converter cannot smoothly reach a zero or unity transfer characteristic without exhibiting some discontinuities 19 and 21 at the extremes of D. This phenomenon occurs due to switching delays in the power MOSFET switch and its control and gate drive circuitry.
As long as the Buck converter's power MOSFET 3 is still switching, tsw is limited to some portion of the clock period T, e.g. 5%<D<95%, essentially due to turn-on and turn-off delay within the MOSFET switch and its control loop. For example, at a 95% duty factor and a 3 MHz clock, the off-time for the high-side MOSFET 3 is only 5% of the 333 nsec period, or just 16 nsec. This means the high side MOSFET 3 must turn off and back in only 16 nsec—too rapidly to regulate over a 95% output-to-input conversion ratio. This minimum off-time problem impacts both synchronous or non-synchronous Buck converters. This problem is further exacerbated in a synchronous DC/DC converter, since no time remains for the synchronous rectifier MOSFET 4 to turn on and then off again and still exhibit BBM operation.
Referring again to graph 15 in FIG. 2A, above some maximum duty factor Dmax, there is not adequate time to maintain switching operation and the converter jumps from Dmax to a 100% duty factor, as shown by discontinuity 21. Above Dmax, the converter turns on MOSFET 3 and leaves it on for the entire period T. The abrupt transition 21 causes a glitch in the output voltage. Thus, at a 100% duty factor, Vout=Vin as shown by line 16 and all regulation is lost as long as the switching is halted.
Synchronous boost converter 10, shown in FIG. 1B includes a low-side power MOSFET 12, a battery-connected inductor 13, a filter capacitor 15, and a “floating” synchronous rectifier MOSFET 14 with a parallel rectifier diode 16. The gates of the MOSFETs 12 and 14 are driven by break-before-make circuitry (not shown) and controlled by a PWM controller 11 in response to voltage feedback VFB from the output voltage Vout across filter capacitor 15. BBM operation is needed to prevent shorting out filter capacitor 15.
The synchronous rectifier MOSFET 14, which may be an N-channel or P-channel MOSFET, is considered to be floating in the sense that neither its source nor drain terminal is permanently connected to any supply rail, i.e. ground or Vbatt. Diode 16 is a P-N diode intrinsic to synchronous rectifier MOSFET 14, regardless whether synchronous rectifier MOSFET 14 is a P-channel or an N-channel device. A Schottky diode may be included in parallel with MOSFET 16, but with series inductance may not operate fast enough to divert current from forward biasing intrinsic diode 16. Diode 17 represents a P-N junction diode intrinsic to N-channel low-side MOSFET 12 and remains reverse biased under normal boost converter operation. Since diode 17 does not conduct under normal operation, it is shown as dotted lines.
If we again define the converter's duty factor D as the time that energy flows from the battery or power source into the DC/DC converter, i.e. during the time that low-side MOSFET switch 12 is on and inductor 13 is being magnetized, then the output to input voltage ratio of a boost converter is proportionate to the inverse of 1 minus its duty factor, i.e.
            V      out              V      in        =            1              1        -        D              ≡          1              1        -                              t            sw                    /          T                    
This relationship for a boost or synchronous boost converter is illustrated by curve 18 in FIG. 2A in graph 15. Notice that the boost converter cannot smoothly reach a unity transfer characteristic without exhibiting some discontinuity at the extremes of D. This phenomenon occurs due to switching delays in the power MOSFET switch and its control and gate drive circuitry.
As long as the boost converter's power MOSFET 12 is still switching, tsw is limited to some portion of the clock period T, e.g. 5%<D<95%, essentially due to turn-on and turn-off delay within the MOSFET 12 and its control loop. For example, at a 5% duty factor and a 3 MHz clock, the on-time for the low-side MOSFET 12 is only 5% of the 333 nsec period, or just 16 nsec. This means the low side MOSFET 12 must turn on and back off in only 16 nsec—too rapidly to regulate below a 5% output-to-input conversion ratio. This minimum on time problem impacts either synchronous or non-synchronous boost converters.
Referring again to graph 15 in FIG. 2A, below some minimum duty factor Dmin, there is not adequate time to maintain switching operation and the converter must jump from Dmin to 0% duty factor as shown by discontinuity 20. Below Dmin, the converter turns on the synchronous rectifier MOSFET 14 and leaves it on for the entire period T. The abrupt transition 20 causes a glitch in the boost converter's output voltage. Moreover, at a 100% duty factor, Vout=Vin as shown by line 16, all regulation is lost as long as the switching is halted.
So in both synchronous Buck converter 1 and synchronous boost converter 10, operation near a unity transfer characteristic, i.e. when Vout≈Vin shown by line 16, is problematic for either the Buck or the boost converter.
The efficiency η of a voltage converter can be given by
  η  =                    P        out                    P        in              =                            I          out                ·                  V          out                                      I          in                ·                  V          in                    
An analysis of inductive switching regulator efficiencies is described in detail the above-referenced application Ser. No. 11/890,818.
Graph 25 of FIG. 2B illustrates examples of typical conversion efficiencies for synchronous Buck and synchronous boost converters as a function of the converter's voltage conversion ratio Vout/Vin. As shown, line 26 represents the unity conversion condition, where Vout=Vin. Conversion ratios less than unity, on the left side of line 26 in graph 25, represent step-down conversion. Efficiency curve 27 represents an example of a Buck converter performing a step-down voltage conversion. Conversion ratios greater than unity, on the right side of line 26, represent step-up conversion. Efficiency curve 28 represents an example of a boost converter performing step-up voltage conversion.
In general, boost regulators exhibit lower efficiencies than Buck regulators for comparable load currents, as illustrated by curves 27 and 28. This is primarily due to the fact that boost regulators exhibit higher peak currents than Buck regulators. This problem is further accentuated for high Vout/Vin voltage conversion ratios, especially for output voltages approaching ten times the input voltage, as illustrated by the decline of curve 28 at higher conversion ratios.
In graph 25, the efficiency of a Buck converter (curve 27) is not shown for conversion ratios below 0.1 or above 0.9 and likewise the efficiency of a boost converter (curve 28) is not shown for conversion ratios below 1.1 or above 10, because these conversion ratios require the converter to operate below a 10% or above a 90% duty factor, an operating condition difficult to achieve, especially at high switching frequencies.
Buck-Boost Switching Converter
The problem of non-isolated DC/DC switching converter operation near unity transfer is especially difficult in applications when the input voltage may vary above or below the desired output voltage. Examples of this application include the output of noisy AC adapters or circuitry which must operate by battery back-up during emergency conditions when a main source of power has failed.
Another scenario where a unity conversion ratio is required occurs when a battery's operating voltage range extends above and below the desired output voltage. For example, the discharge characteristic of a Lilon battery starts at 4.2V at full charge, initially decays rapidly to around 3.6V, then decays slowly from to 3.4V, and finally drops quickly to its cutoff at or below 3V. In the event that a DC/DC converter is needed to produce a well-regulated 3.3V output during this entire period, a sub-unity conversion ratio of (3.3V/4.2V), i.e. a ratio of 0.79, is needed at the outset, indicating a Buck converter is required. At the battery's end-of-life, the required conversion ratio exceeds unity, becoming 3.3V/3V, i.e. a conversion ratio of 1.1, and requires a boost converter. Such an application demanding both step-up and step-down conversion requires a Buck-boost, or up-down converter.
In the case where the user wants to avoid the complexities of up-down conversion, one possible approach is to use only a Buck converter and give up some battery life by cutting of the battery early, e.g. at 3.3V. In practice, however considering battery manufacturing variations and regulator drop-out and duty factor limitations, too much battery life is sacrificed to rely on a Buck-only regulator solution.
If up-down conversion cannot be avoided, a Buck-boost converter can easily be derived from combining synchronous Buck and boost converters into a merged circuit. In FIG. 3A, for example, a cascade Buck-boost converter 35 contains a synchronous Buck converter comprising a P-channel or N-channel MOSFET 36, an inductor 38A, an N-channel synchronous rectifier MOSFET 37 with an intrinsic rectifier diode 39, and a capacitor 44, which is used to power a synchronous boost converter comprising a low-side N-channel MOSFET 40, an inductor 38B, a synchronous rectifier MOSFET 41 with an intrinsic rectifier diode 42, and a filter capacitor 43. Buck-boost converter 35 first steps down the input voltage Vbatt to an intermediate voltage lower than the desired output, then steps the intermediate voltage up to produce Vout.
FIG. 3B conversely illustrates a cascade boost-Buck converter 45 that contains a synchronous boost converter comprising a low-side N-channel MOSFET 46, an inductor 47, an N-channel or P-channel synchronous rectifier MOSFET 48A with an intrinsic diode 49, and a capacitor 54, which is used to power a synchronous Buck converter comprising a MOSFET 48B, an inductor 52, an N-channel synchronous rectifier MOSFET 50 with an intrinsic rectifier diode 51, and a filter capacitor 53. Buck-boost converter 45 drives a load (not shown). In this approach, the input voltage Vbatt is first stepped-up to an intermediate voltage higher than the desired output, then back down to produce Vout.
The overall efficiency of either Buck-boost regulator 35 or boost-Buck regulator 45 is given by the product of the boost converter's efficiency ηboost multiplied by the Buck converter's efficiency ηBuck. Mathematically this can be represented as ηcascade=ηBuck·ηboost. Even if both converters are 85% efficient, the efficiency of the cascade Buck-boost or boost-Buck converter reaches an overall efficiency of only about 70%, significantly lower than the typical efficiency of either a Buck converter or a boost converter alone. The overall power loss in a cascaded Buck-boost or boost-Buck cascade is greater than the power loss in either a synchronous Buck or synchronous boost converter alone, because there are more transistors in series between input and output terminals, and because all the transistors are switching all the time.
As shown in FIG. 3B, boost-Buck converter 45 includes series-connected MOSFETs 48A and 48B with an intermediate capacitor 54. Since in steady-state, the current in series connected MOSFETs must be equal, MOSFET 48B is redundant and can be eliminated without impacting circuit operation. Even so, boost-Buck converter 45 requires two inductors 47 and 52, a characteristic highly undesirable from a user's point-of-view.
Similarly, as shown in FIG. 3A, Buck-boost converter 35 includes inductors 38A and 38B with intermediate capacitor 44. Since in steady state the current in inductors 38A and 38B is the same, inductor 38B is redundant and may be eliminated without changing the function of the circuit. In fact, capacitor 44 may also be eliminated without significantly altering the operation of Buck-boost converter.
The resulting simplified prior-art Buck-boost converter 55 is illustrated in FIG. 3C. Buck-boost converter 55 comprises a single-inductor 59; four MOSFETs 57, 56, 60, and 61; diodes 58 and 62 and a filter capacitor 63. The PWM controller and break-before-make and gate buffer circuits are not shown. Depending on its terminal conditions, such a converter can operate in three distinct modes, Buck, boost, and Buck-boost.
In FIG. 3D, equivalent circuit diagram 65 represents the operation of Buck-boost converter 55 as a Buck converter where MOSFETs 57 and 56 are switched out-of-phase under PWM control while MOSFET 61 remains turned-on, represented as a resistance 67, and MOSFET 60 is biased off, shown as an open circuit 66. The overall power loss in Buck-boost converter 55 operated as a Buck converter is greater than that in an equivalent synchronous Buck converter because of the conduction loss in MOSFET 61, i.e. power lost continuously in resistance 67. As a result of this increased power loss, Buck-boost converter 55 operating in its Buck mode has a lower efficiency than conventional Buck converter 1 shown in FIG. 1A.
In FIG. 3E, equivalent circuit diagram 70 represents the operation of Buck-boost converter 55 as a boost converter where MOSFETs 60 and 61 are switched out-of-phase under PWM control while MOSFET 57 remains turned-on, represented as a resistance 71, and MOSFET 56 is biased off, shown as an open circuit 72. The overall power loss in Buck-boost converter 55 operated as a boost converter is greater than that in an equivalent synchronous boost converter because of the conduction loss in MOSFET 57, i.e. power lost continuously in resistance 71. As a result of this increased power loss, Buck-boost converter 55 operating in its boost mode has a lower efficiency than conventional boost converter 10 shown in FIG. 1B.
The loss of efficiency using Buck-boost converter 55 is illustrated in FIG. 4 in the plot of efficiency η for various output-to-input voltage conversion ratios Vout/Vin. For convenience, the efficiency of conventional Buck and boost converters (similar to curves 27 and 28 in FIG. 2B) is illustrated by curves 81 and 82, respectively.
Curve 83 illustrates the efficiency of Buck-boost converter 55 operating in Buck-only mode, as shown in equivalent circuit diagram 65 (FIG. 3D). Because of series resistance 67 associated with on-state MOSFET 61, the efficiency of a Buck-boost converter in the Buck mode (curve 83) is lower than that of the efficiency of a simple Buck (curve 81). This loss of efficiency can range from a few percent to over 10%, depending on operating conditions. Curve 85 illustrates the efficiency of Buck-boost converter 55 operating in full Buck-boost mode where all four switches are switching constantly, and as a result exhibits even greater losses and poorer efficiency than the same Buck-boost converter operating in Buck mode (curve 83).
Curve 84 illustrates the efficiency of Buck-boost converter 55 operating in boost-only mode, shown in equivalent circuit diagram 70 (FIG. 3E). Because of series resistance 71 associated with on-state MOSFET 57, the efficiency of a Buck-boost converter in the boost-only mode (curve 84) is lower than the efficiency of a simple boost converter (curve 82). This loss of efficiency can range from a few percent to over 10%, depending on operating conditions. Curve 86 illustrates the efficiency of Buck-boost converter 55 operating in full Buck-boost mode where all four switches are switching constantly, and as a result exhibits even greater losses and poorer efficiency than the same Buck-boost converter operating in boost mode (curve 84).
Near a unity conversion ratio, where the output voltage is slightly above or below its input (i.e. where Vout≈Vin) Buck-boost converter 55 must operate in the Buck-boost mode with all four MOSFETs switching constantly. The resulting efficiency (curve 87) can be 10% to 20% lower than the efficiency of a conventional Buck or boost converter (curves 81 and 82).
Thus, the efficiency penalty of using a Buck-boost converter in order to operate over a wide range of voltage conversion ratios is substantial. Moreover, the converter must change its operating mode whenever operating near unity voltage conversion ratios.
Charge Pump Converters
An alternative to the switched-inductor converter is a charge pump, a voltage conversion circuit using only switches and capacitors to perform voltage translation through repeated charge redistribution, i.e. the continuous charging and discharging of a capacitor network driven by a clock or oscillator.
The advantage of a charge pump is that at specific voltage conversion ratios, it can exhibit extremely high conversion efficiencies approaching 100%. The disadvantage is that it can only efficiently generate an output voltage that is a predetermined multiple of the input voltage, based on the number of flying capacitors used in its converter circuit. When used to generate voltages other than a select multiple of the input voltage, charge pumps exhibit low efficiencies.
An example of a common charge pump is illustrated by charge pump 90 in FIG. 5A where a single flying capacitor 93 is employed as a “doubler”, i.e. to double the battery's input voltage. Charge pump 90 comprises MOSFETs 92, 91, 94 and 95, configured in an arrangement similar to an H-bridge except that one terminal of the H-bridge, the source of MOSFET 95, is connected to the output terminal of charge pump 90 and to reservoir capacitor 96 rather than to ground.
Operation of charge pump 90 involves repeatedly charging and discharging flying capacitor 93. During the charging phase, diagonal MOSFETs 94 and 91 are closed, charging capacitor 93 to the voltage Vbatt while MOSFETs 92 and 95 remain open. Thereafter, in the charge transfer phase, MOSFETs 94 and 91 are opened, MOSFETs 92 and 95 are closed, and energy is transferred from the flying capacitor 93 to the output reservoir capacitor 96, pumping the output voltage VCP to a value twice the battery voltage Vbatt.
The purpose of the switch network is essentially to place the flying capacitor in parallel with the battery during the charging phase and in series, i.e. stacked on top of the battery's positive terminal, during the discharging phase, as illustrated by equivalent circuit 100 in FIG. 5B, where voltage source 101 represents the battery input and capacitor 102 charged to Vbatt represents the flying capacitor 93. By “stacking” the charged flying capacitor 93 atop the battery, the output voltage of the charge pump is the sum of the voltages, hence doubling the voltage input. The cycle then repeats with another charging phase.
FIG. 5C illustrates a charge pump 110 utilizing two flying capacitors 114 and 115 and a network of seven MOSFETs 111, 112, 113, 116, 117, 118 and 119. The purpose of the network initially is to charge the capacitors 114 and 115 in series, with each of capacitors charged to half the battery voltage, i.e. Vbatt/2. During charging, MOSFETs 111, 112 and 113 are on and MOSFETs 116, 117, 118 and 119 are off. After charging, the charged capacitors 114 and 115 are connected in parallel, and connected to the positive terminal of the battery. This connection is accomplished by turning on MOSFETs 116, 117, 118 and 119. The resulting output voltage, as shown in the equivalent circuit 121 of FIG. 5D is equal Vbatt+Vbatt/2, for an output voltage of 1.5Vbatt. As shown, battery voltage source 124 and the parallel combination of capacitors 122 and 123 are stacked atop one another. Because the output voltage is 1.5 times the input voltage this type of charge pump is sometimes referred to as a “fractional” charge pump.
Actually, many charge pump topologies are possible, but most use only one or two flying capacitors. A single flying capacitor charge pump is only capable of efficiently delivering power at twice its input, or alternatively if the capacitor is connected to the negative terminal of the battery to produce a mirror-image negative voltage of the battery, i.e. −Vbatt, also known as an inverter. The inverting case is illustrated in equivalent circuit 130 of FIG. 5E, where battery 131 is used to charge capacitor 132 to a voltage below ground, i.e. referenced to the negative terminal of battery 131. Two-transistor fractional charge pumps may be used to produce an output voltage equal to one-half the input voltage, as shown in equivalent circuit 135 of FIG. 5F where capacitors 137 and 138, after being charged to one-half of the battery voltage 136, are then referenced to the negative battery potential (ground) to generate a positive potential equal to +0.5Vbatt. Alternatively, the positive sides of capacitors could be connected to ground to generate an inverted potential equal to −0.5Vbatt.
The problem with charge pump converters is they operate efficiently only at specific conversion multiples determined by the number of flying capacitors. In other words, they are not voltage regulators. Specifically, as a desired load voltage Vout deviates from the voltage VCP that the capacitor network produces, the charge pump cannot adapt. To bridge the voltage-differential between the charge pump's output voltage VCP and the desired output voltage Vout requires a resistor or current source, and the voltage across that lossy element results in lost power and reduced efficiency. An analysis of charge pump efficiencies is provided in the above-referenced application Ser. No. 11/890,818.
The efficiency equation for single-mode charge pumps is illustrated graphically in FIG. 6A for various multipliers, including a doubler (curve 151), an inverter (curve 152), and fractional charge pumps (curves 153, 154 and 155). Curve 156 represents the efficiency of a charge pump designed to generate an output voltage equal to its input voltage, identical to a linear regulator's maximum theoretical efficiency, i.e. assuming no quiescent operating current. In each case, the efficiency of the charge pump increases as the ratio of the output voltage to the input voltage approaches an integral multiple of ±½Vbatt. Above that voltage ratio, the charge pump is not capable of operating, and a different capacitor multiplier, i.e. a different operating mode, must be employed.
Each curve shown in graph 150 represents a specific charge pump circuit, e.g. including those shown in FIGS. 5A-5F. Unless a load operates at an exact half-volt integral multiple of the input voltage, however, the efficiency of a charge pump converter using one or two capacitors will suffer. This behavior is especially problematic for battery-powered products, since the battery voltage may change markedly as the cell discharges. In the case of Lilon batteries, for example, the voltage can decay more than 1V during discharge, representing a 25% change. Even if the peak efficiency may be high at one specific operating condition and battery voltage, the overall efficiency of the converter averaged over the battery discharge curve is poor. Weighted average efficiencies can be lower than 60% using a single-mode charge pump.
One way to improve the average efficiency of the converter is to switch modes between conversion ratios of 1X, 1.5X and 2X automatically within one circuit. This feature is particularly useful to supply a fixed voltage over a wide input range. The efficiency of a mode-changing charge pump is illustrated in FIG. 6B, where as the battery decays the tri-mode converter circuit switches from a 1X-battery-direct mode having an efficiency shown by curve 163, to a 1.5X-fractional-mode having an efficiency shown by curve 162, and then to 2X-doubler-mode having an efficiency shown by curve 161. By switching modes in this zigzag pattern, the efficiency of the charge pump converter is improved because the output is not pumped to an excessively high value compared to the load.
Unfortunately, conditions still exist where the efficiency suffers substantially. The mode transitions exhibit dramatic shifts in efficiency at a conversion ratio of one (curve 163), and again at a conversion ratio of 1.5 (curve 162). The mode transitions may also result in sudden current and voltage discontinuities, or produce instability or noise. To determine what conversion ratio, is required graph 160 also includes curves 166, 165, and 164, relating the input voltage range and conversion ratios required to produce output voltages of 3V, 3.5V and 4V, respectively.
Specifically, the charge pump converter in 1.5X mode does not perform well for conditions slightly above a unity conversion ratio, unfortunately manifesting efficiencies that are even lower than the efficiency of an inductive Buck-boost converter.
Dropout in Prior Art Regulators
Whenever the input and the output voltages of a voltage converter approach a range of several hundred milli-volts of each other, e.g. Vout≈Vin±200 mV, the quality of the converter's regulating ability suffers. Loss of regulation quality may be manifest in several ways, either by a one-time or repeated glitch or discontinuity in output voltage, by increased ripple, or by complete loss of regulation within some narrow voltage band. The phenomenon of degraded regulation whenever Vout approaches Vin is referred to as “dropout”, meaning the converter “drops out” of regulation.
The Buck converter 1 of FIG. 1A and the boost converter 10 of FIG. 1B both momentarily lose regulation as their switching duty factor jumps from Dmax or Dmin to 100% and they completely lose regulation while D=100% since the input is essentially resistively connected to the output during the dropout condition.
While the Buck-boost converter doesn't really exhibit permanent dropout, it can easily suffer a voltage glitch during mode transitions whenever the converter mode switches from a Buck converter into its Buck mode into its Buck-boost mode, or when switching from Buck-boost mode to boost mode. Mode transitions occur whenever the converter changes from a circuit having two power devices switching into one where four devices are switching, or vice versa.
To avoid the mode-switching problem, a Buck boost converter can be run continuously in Buck-boost mode, with all four power MOSFETs switching continuously, but then its efficiency is degraded under all input-output conditions and conversion ratios.
As stated previously, the charge pump is incapable of regulating voltage without the use of a series connected linear regulator to provide the regulation function. Unfortunately, it is well known phenomenon that all linear regulators exhibit loss of regulation, i.e. dropout, whenever the ΔV across the linear regulator's input and output terminals becomes too small. In essence, dropout occurs in a linear regulator because the loop gain of the amplifier performing regulation drops precipitously as its transistor pass element changes from behaving as a current source to behaving as a variable resistor. If the pass element is a bipolar transistor, the loss of gain occurs at small values of VCE as the device transitions from its active operating region into saturation. In many bipolar linear regulators, this dropout condition occurs at more than 400 mV.
In so-called “low dropout” linear regulators, or “LDOs”, a MOSFET capable of operating as a current source at a lower ΔV is substituted for the bipolar pass element, but the linear regulator still drops out at 200 to 300 mV as the power MOSFET pass element transitions from its saturation, i.e. constant current, region, into its linear, i.e. resistive, region of operation.
In conclusion, all prior-art non-isolated high-efficiency converters exhibit dropout at voltage conversion ratios approaching unity. Mode switching, loss of regulation and dropout can be avoided, but only by sacrificing efficiency. Isolated converters such as the flyback and forward converter are able to operate at high efficiencies near unity conversion without the need to switch modes, but their use of physically-large tapped inductors, coupled inductors, and transformers precludes their application in most portable products.
Summary of Prior-Art Down-Up Converters
In conclusion, no existing charge pump converter, Buck-boost switching regulator or other inductive switching regulator is able to both step-up and step-down DC voltages efficiently, especially for conversion ratios near unity where Vin≈Vout. What is needed is an up-down converter that is efficient over a wide range of input and output voltages, and that does not need to change its operating mode as it approaches or operates near unity voltage conversion ratios. Furthermore the converter should be free from dropout problems, maintaining high-quality regulation even while biased with an output voltage within 200 mV of its input voltage, i.e. where Vout≈Vin±200 mV.